1. Field of the Invention
The present invention relates to the field of current-isolated indirect-feedback instrumentation amplifiers.
2. Prior Art
Instrumentation amplifiers are considered closed loop gain blocks with high gain accuracy and excellent common mode rejection ratio (CMRR) performance. Non-sampled instrumentation amplifiers suitable for precision analog integrated circuits on silicon chips can be divided into two main categories.
1. Bridge type instrumentation amplifiers, and
2. Current isolation instrumentation amplifiers.
The bridge type instrumentation amplifier is the most widely used instrumentation amplifier, which is built around an opamp (operational amplifier) and four resistors in a bridge configuration. A typical bridge type instrumentation amplifier stage is shown in FIG. 1a. As shown in FIG. 1b, two non-inverting buffers with a few resistors often precede this stage for increased input impedance and gain variability using an external resistor. Nevertheless, the gain accuracy and common mode rejection ratio of such circuits are still very much dependant on the bridge section, sometimes called the difference amplifier.
Although bridge type instrumentation amplifiers are known for excellent linearity and gain accuracy, they often suffer from the degraded common mode rejection ratio at lower gains, besides the difficulty with lower rail sensing (depending on the opamp characteristics). For example, in FIG. 1b the output of opamps of A1 and A2 cannot reach the GND, thereby prohibiting the rail sensing of the instrumentation amplifier. They also require large integrated circuit area (for incorporating accurate opamps) which makes them unsuitable for packaging in small commercial packages, such as an SOT23 package.
The common mode rejection ratio of bridge type instrumentation amplifiers is characterized by the following equation:
1/CMRR=1/CMRRopamp+4xcex5/(1+G)
Were:
CMRRopamp is the CMRR of the operational buffer used to build the difference amplifier
xcex5 is the resistor mismatch of the bridge
G is the open loop gain of the amplifier
From the above, it is quite evident that even with a perfect common mode rejection ratio for the opamp, the mismatch of the bridge, especially at lower gains, can degrade the overall common mode rejection ratio to poor values unfit for most applications. For example, with a 0.04% mismatch, the overall common mode rejection ratio can only reach 62 dB of rejection.
The second class of instrumentation amplifiers is based on the conversion of the input differential voltage to a current through a transconductance amplifier, and then the conversion of the current back to an output voltage in order to isolate the input stage and its common mode voltage from the rest of the circuitry (see FIG. 2).
Indirect current feedback instrumentation amplifiers, as shown in FIG. 3, are based on the conversion of the input differential voltage VIN and a portion (VFB) of the output voltage VOUT (the voltage across resistor R1 in the series combination of resistors R1 and R2), to two differential currents by transconductance amplifiers GM1 and GM2. These currents are then subtracted from each other and fed to a high gain transimpedance amplifier A at the output to close the loop.
A more detailed circuit for such an instrumentation amplifier is shown in FIG. 4. Here, differential amplifiers GM1 and GM2 are comprised of transistors P1 and P2 and resistors R11 and R12, and transistors P3 and P4 and resistors R21 and R22, respectively, together with current sources I.
Differential transconductance amplifier GM1 at the input and differential transconductance amplifier GM2 at the output, together with the transimpedance loop amplifier A (for differential current to voltage conversion) comprise the indirect current feedback architecture.
For such a configuration, assuming a high gain for the loop amplifier A, the output currents of the two transconductance blocks algebraically sum to zero. Thus the gain equation is:
Gain=(GM1/GM2)(1+R2/R1)
One limitation of such an arrangement, which is somewhat inherent to the very fundamental operation of the differential pairs with relatively large signals at their inputs, is the change of transconductance GM as a function of the common mode of the differential inputs. In that regard, note in FIG. 4 that the positive input of GM2 is referenced to ground, or alternatively referenced to a reference voltage, while each of the differential inputs VIN of GM1 is user controlled.
The transconductance of a differential pair can be defined as:
GM=∂Iod/∂Vid=f(Vid, Vicm, Vbd, Vbcm)
Where:
Iod=differential output current of the differential pair
Vid=input differential voltage
Vicm=input common mode voltage
Vbd=differential body voltage
Vbcm=common mode body voltage
(see FIGS. 5a and 5b)
The change in GM due to Vicm is caused by the dependency of GM on Vds for a MOS transistor. For large common mode swings, the nonlinearity in GM can be high, which in turn can be translated into a gain error, especially when intrinsic gains of the input devices are low, as in the case of a MOS differential pair. (The intrinsic gain is the product of GM and ro, where ro is the output resistance of the transistor which is equal to the Early voltage (VA) divided by the drain current ID (ro=VA/ID). In FIG. 4, the differences in GM of the two differential pairs due to differences in the input common mode voltages of the two differential pairs will be such that the non-linearity of one pair is no longer canceled out by the non-linearity of the other pair. This gives rise to excessive non-linearity error.
Another limitation of the circuit in FIG. 4 is its low common mode rejection ratio CMRR, especially for MOS input devices. It can be shown that for any differential stage, the overall CMRR can be approximated as:
1/CMRR=(1/xcexc)(xcex94xcexc/xcexc)+(1/CMRR associated with the tail current source, here ignored)
Where:
1/xcexc=1/((GM)(ro))=1/(average intrinsic gain of the input transistors), called the isolation factor
xcex94xcexc/xcexc=normalized difference between the intrinsic gains of input transistors, called the balancing factor
For bipolar devices having an isolation factor on the order of 1/xcexc=1/1000 and a balancing factor of xcex94xcexc/xcexc=2/100, a common mode rejection ratio of 94 dB is achievable. However, in MOS devices of 1/xcexc=1/50 and xcex94xcexc/xcexc=5/100, the common mode rejection ratio is 60 dB, or less if the isolation or balancing factors tend to have even more degraded values.
On the other hand, MOS devices at the input of instrumentation amplifiers have the benefit of very high input impedance, in addition to providing the luxury of controlling the GM of the input stage by yet another parameter (W/L of the MOS devices) in addition to the input tail current. Using such a MOS differential pair for GM1 and GM2 in FIG. 4 results in lower overall common mode rejection ratio, and indirectly lower again accuracies through secondary effects of the input common mode voltage on GM.
Also known in the prior art is the constant GM bias circuit (FIG. 6) and the observation made by Roel Wassenaar. Roel Wassenaar made the observation that any differential pair with its transistors of the same length (L), same current density (I/W), where W is the width, and same type (bipolar or MOS of similar type) as the transistors in the bias circuit itself, and with the tail current of the differential pair fed from the bias circuit, will have the same GM as the bias circuitry itself.
Writing the translinear loop of VGSP1, VGSP2 and R:       I    =                  K        1            ⁢              1                  R          2                    ⁢              xe2x80x83            ⁢      strong      ⁢              xe2x80x83            ⁢      inversion            I    =                  K        2            ⁢              1        R            ⁢              xe2x80x83            ⁢      weak      ⁢              xe2x80x83            ⁢      inversion      
Where K1 and K2 are process, size (W/L) and fold (N) dependant. Since GM is proportional to {square root over (I)} in strong inversion and proportional to I itself in weak inversion, then:       GM    ≅                  K        3            ⁢              1        R            ⁢              xe2x80x83            ⁢      strong      ⁢              xe2x80x83            ⁢      inversion      ⁢              xe2x80x83            ⁢      or            GM    ≅                  K        4            ⁢              1        R            ⁢              xe2x80x83            ⁢      weak      ⁢              xe2x80x83            ⁢      inversion      
If transistor P4 is supplying the tail current of a differential pair with transistors of the same type, same length and same current density as transistors in the bias circuitry itself, then the GM of the differential pair is similar to the GM of the bias circuit.
The GM-controlled current-isolated indirect-feedback instrumentation amplifiers of the present invention use two transconductance models to set the I/V characteristic of the two GM blocks in such instrumentation amplifiers. Any mismatch between the two transconductance blocks will be seen as a gain accuracy error. Moreover, since the common mode input voltage can change the transconductances, any mismatch between the transconductances GM1 and GM2 will no longer cancel each other as they change through common mode changes (non-linearity error). Also, for large differential signal swings, the I/V transfer function transconductance blocks tends to flatten, which will cause some non-linearity error due to GM reduction inherent to the differential pairs.
By adding two differential GM models, the transconductors of those two blocks (GM1 and GM2) will match the transconductance of their respective models. The models have the same transconductances equal to some reciprocal of a resistor over the input voltage range, process and temperature. Therefore the two GM models will have equivalent transconductances which will track over the input voltage range (common mode and differential mode), process and temperature.
The benefits of using this new technique include:
1. An increase in gain accuracy through better matching of GM1 and GM2, which are proportional to 1/RAB and track over temperature, process and voltage.
2. Better gain linearity of the GM blocks as the tail currents of GM1 and GM2 are regulated with the differential input voltage. Also, the GM circuits are now regulated independent of their common mode voltage.
3. An increase in the common mode rejection ratio due to the regulation of the tail currents of the GM circuits.